Sensing and feedback in a current mode control voltage regulator

ABSTRACT

The disclosed embodiments of voltage regulators incorporate a current mode control architecture. In one embodiment, a comparator mechanism triggers a transition in a power switch when the error in the regulated output voltage is equal to a proportionally scaled value of current provided at an output filter. The voltage regulator includes a power switch having an input and an output. The power switch is configured to provide a first voltage during a first conduction period and a second voltage during a second conduction period. An output filter is coupled between the power switch output and an output terminal to be coupled to a load. A comparator mechanism has a reference input coupled to a reference voltage, a feedback input coupled to sense a feedback voltage at the output filter, a current sensing input coupled to sense a current sensing voltage corresponding to a current provided to the output filter, and an output in communication with the power switch input. The comparator mechanism is configured to trigger responsive to a difference between the feedback voltage and the reference voltage equaling the current sensing voltage. The triggering causes a transition of the power switch from the second conduction period to the first conduction period.

PRIORITY DATA

The present application claims priority to co-pending and commonlyassigned U.S. Provisional Patent Application No. 61/368,131, titledSENSING AND FEEDBACK IN A CURRENT MODE CONTROL VOLTAGE REGULATOR, byTournatory, et al., filed on Jul. 27, 2010, which is hereby incorporatedby reference in its entirety and for all purposes.

BACKGROUND

The present invention relates generally to voltage regulators, and moreparticularly to the architecture and control mechanisms of switchingvoltage regulators.

Voltage regulators, such as direct current (DC) to DC converters, areused to provide stable voltage sources for electronic devices andsystems. The general purpose of a voltage regulator is to convert asource voltage, such as the voltage of an alternating current (AC) or DCpower source, into the operating DC voltage of an electronic device.Efficient DC to DC converters are used for battery management in lowpower devices, such as laptop notebooks and cellular phones.

Switching voltage regulators, often referred to as switching regulators,are a type of DC to DC converter that convert one DC voltage to anotherDC voltage with high efficiency. A switching regulator generates anoutput voltage by converting an input DC voltage into a high frequencyvoltage, and filtering the high frequency voltage to produce the outputDC voltage.

Conventional switching regulators typically include a switch foralternately coupling and decoupling an unregulated input DC voltagesource, such as a battery, to a load, such as an integrated circuit. Anoutput filter, typically including an inductor and a capacitor, iscoupled between the switch and the load to filter the output of theswitch and thus provide the output DC voltage. Power is transmittedthrough the switch and into the output filter in the form of discretecurrent pulses. The switching regulator operates on the principle ofstoring energy in the inductor during one portion of a cycle and thentransferring the stored energy to the capacitor in the next portion ofthe cycle. The output filter converts the current pulses into a steadyload current so that the voltage across the load is regulated.

SUMMARY

According to one aspect of the present invention, an embodiment of acomparator mechanism triggers a transition in a power switch when theerror in the regulated output voltage is equal to a proportionallyscaled value of current provided at an output filter.

According to one aspect of the invention, a voltage regulator includes apower switch having an input and an output. The power switch isconfigured to provide a first voltage during a first conduction periodand a second voltage during a second conduction period. An output filteris coupled between the power switch output and an output terminal to becoupled to a load. A comparator mechanism has a reference input coupledto a reference voltage, a feedback input coupled to sense a feedbackvoltage at the output filter, a current sensing input coupled to sense acurrent sensing voltage corresponding to a current provided to theoutput filter, and an output in communication with the power switchinput. The comparator mechanism is configured to trigger responsive to adifference between the feedback voltage and the reference voltageequaling the current sensing voltage. The triggering causes a transitionof the power switch from the second conduction period to the firstconduction period.

In one implementation, the current sensing voltage represents thecurrent provided to the output filter multiplied by an effectiveresistance of a component of the power switch, such as a transistor. Inanother implementation, the current sensing voltage represents thecurrent provided to the output filter multiplied by an effectiveresistance of a component of the output filter, such as an inductor. Inanother implementation, the current sensing voltage represents thecurrent provided to the output filter multiplied by an effectiveresistance of a resistor coupled at the output filter, such as anexternal sense resistor.

According to another aspect of the present invention, the comparatormechanism comprises a sampling circuit coupled to sample the referencevoltage during the first conduction period. A multiplexer has thereference input and the feedback input as inputs. A capacitor has aninput coupled to an output of the multiplexer. A switch is coupled to anoutput of the capacitor. A comparator has a first input coupled to thecapacitor output, the current sensing input as a second input, and anoutput representing the comparator mechanism output. The output of thecomparator is in communication with a select line of the multiplexer anda control of the switch.

According to another aspect of the present invention, a state registerhas an input coupled to the comparator output and an output coupled to:the power switch input, the multiplexer select line, and the switchcontrol. A switching control unit is coupled to a control input of thestate register. The state register is capable of: initiating thetransition of the power switch to the first conduction period responsiveto the comparator mechanism output, and initiating a transition of thepower switch to the second conduction period responsive to the switchingcontrol unit. This includes causing the multiplexer to output thereference voltage during the first conduction period and the feedbackvoltage during the second conduction period, and causing the switch tobe closed during the first conduction period and open during the secondconduction period.

In one implementation, the switching control unit comprises a timer, forinstance, having a fixed on time or a variable on time. In anotherimplementation, the switching control unit comprises a clock.

According to another aspect of the present invention, the comparatormechanism comprises a continuous tracking circuit coupled to sense thereference voltage. The continuous tracking circuit of the comparatormechanism comprises a first current sensing voltage input and a secondcurrent sensing voltage input coupled to sense a differential voltagerepresenting the current sensing voltage. In one implementation, thefirst and second current sensing voltage inputs each comprise acapacitor, and the reference input and the feedback input each comprisea capacitor. A comparator has a first input coupled to the referenceinput and the feedback input, and a second input coupled to the firstcurrent sensing voltage input and the second current sensing voltageinput, and an output representing the comparator mechanism output.

According to another aspect of the present invention, an integratormechanism is incorporated. The integrator mechanism is coupled to sensethe feedback voltage and the reference voltage and has an output coupledto one of the comparator mechanism inputs. The integrator mechanism isconfigured to determine a difference between the feedback voltage andthe reference voltage and output an adjusted signal, based on thedetermined difference, to the comparator mechanism.

Another aspect of the present invention relates to a voltage regulationmethod. A first voltage is provided during a first conduction period,and a second voltage is provided during a second conduction period. Acurrent sensing voltage is sensed corresponding to a current provided tothe output filter. A trigger event is determined when a differencebetween a feedback voltage at the output filter and a reference voltageequals the current sensing voltage. The trigger event causes atransition from the second conduction period to the first conductionperiod.

A further understanding of the nature and advantages of the presentinvention may be realized by reference to the remaining portions of thespecification and the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The included drawings are for illustrative purposes and serve only toprovide examples of possible structures and process steps for thedisclosed inventive devices, circuits, components, systems, and methods.These drawings in no way limit any changes in form and detail that maybe made to the invention by one skilled in the art without departingfrom the spirit and scope of the present invention.

FIG. 1 is a circuit diagram of the main components of a voltageregulator 100, according to an embodiment of the invention.

FIG. 2 is a circuit diagram of an alternative embodiment of a comparatormechanism 200 of voltage regulator 100 and associated input signals,according to an embodiment of the invention.

FIG. 3A is a circuit diagram of the main components of a voltageregulator 300A, according to an embodiment of the invention.

FIG. 3B is a circuit diagram of the main components of a voltageregulator 300B, according to an embodiment of the invention.

FIG. 4 is a circuit diagram of the main components of voltage regulator100, configured according to another embodiment of the invention.

FIG. 5 is a circuit diagram of a resistance adjustment device 400connected to a current sensing component of a voltage regulator,according to an embodiment of the invention.

FIG. 6 is a circuit diagram of a circuit 600 as one implementation ofresistance adjustment device 400, according to an embodiment of theinvention.

DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS

Reference will now be made in detail to specific embodiments of theinvention including the best modes contemplated by the inventors forcarrying out the invention. Examples of these specific embodiments areillustrated in the accompanying drawings. While the invention isdescribed in conjunction with these specific embodiments, it will beunderstood that it is not intended to limit the invention to thedescribed embodiments. On the contrary, it is intended to coveralternatives, modifications, and equivalents as may be included withinthe spirit and scope of the invention as defined by the appended claims.In the following description, specific details are set forth in order toprovide a thorough understanding of the present invention. The presentinvention may be practiced without some or all of these specificdetails. In addition, well known features may not have been described indetail to avoid unnecessarily obscuring the invention.

The disclosed embodiments of the invention relate to and compriseswitching voltage regulators and specific mechanisms to facilitatevoltage conversion. Embodiments of the invention are generally describedherein in relation to a buck regulator, which converts an input (DC)voltage to a lower output voltage of the same polarity. It should beunderstood that embodiments of the present invention also encompassboost regulators, in which the output voltage is higher than the inputvoltage, and buck-boost regulators, which incorporate elements of bothbuck regulators and boost regulators and are capable of reversing thepolarity of the input voltage.

Voltage regulators constructed in accordance with some embodiments ofthe present invention incorporate a power switch to control the flow ofcurrent into the output filter. The power switch is generally configuredto alternatively couple the output filter to a supply voltage, that is,an input voltage source to be regulated, and another voltage, which isoften ground.

In one embodiment, the power switch incorporates a “high side” switchcomponent, such as a transistor, and a “low side” switch component, suchas a transistor or a diode. In one embodiment, the high side switchcomponent is coupled to the supply voltage, while the “low side” switchcomponent is connected to ground. As used herein, an individual highside or low side switch component can be referred to as a high side orlow side “switch.” In this embodiment, the high side switch selectivelycouples the output filter to the supply voltage, while the low sideswitch selectively couples the output filter to ground. The ratio oftime spent with the “high side” switch enabled versus the “low side”switch enabled determines the output voltage developed, for instance, byan LC output filter coupled at the output of the power switch.

A power switch transistor can be implemented as a field effecttransistor (“FET”), such as a metal oxide semiconductor field effecttransistor (“MOSFET”), as illustrated in FIG. 1. The FETs can bep-channel or n-channel, depending on the desired implementation. In analternative embodiment, a different type of transistor is used, such asa junction gate field effect transistor (“JFET”).

FIG. 1 illustrates the major components of a buck voltage regulator 100,constructed according to one embodiment of the invention. In FIG. 1,selected feedback signals are supplied to a comparator mechanism 112.The voltage regulator 100 is constructed with a high side FET (switch)104 and a low side FET (switch) 108 as described above. The high sideswitch 104 is coupled to an input voltage source to be regulated(“V_(DD)”) 128, while the low side switch 108 is coupled to Ground(“Gnd”) at node 136. A switching node Vx 132 is situated at the outputof the power switch comprising high side switch 104 and low side switch108. In particular, node Vx is located between the high side FET 104 andlow side FET 108, in this implementation, between the source of FET 104and the drain of FET 108. The Vx node leads to inductor 148 and outputcapacitor 152 which are considered at least part of an output filter.The output filter is generally coupled to a load (not shown) such as anintegrated circuit.

In FIG. 1, the current delivered to the inductor 148 of the outputfilter through node Vx 132, referred to herein as “I_(L),” ramps up anddown in sequence with the switching between high side FET 104 and lowside FET 108 of the power switch. In particular, when the high side FET104 is turned on, I_(L) ramps up, and when the low side FET 108 isturned on, I_(L) ramps down.

The comparator mechanism 112 directly monitors particular signals ofinterest to determine when to execute the low to high side FETtransition at the power switch. In one embodiment, comparator mechanism112 comprises a comparator 114 and a multiplexer 116, and alsoincorporates an ancillary switch 118 and a capacitor 119, connected asshown in FIG. 1. The comparator 114 is preferably a high-speedcomparator. The comparator mechanism 112 has three input terminals, twoof which are voltage inputs, and the third being a current sensinginput. The first voltage input terminal of comparator mechanism 112,also an input to multiplexer 116, is connected to a reference voltage(“V_(REF)”) 144, and the second voltage input terminal of comparatormechanism 112, also the second input of multiplexer 116, is connected tothe feedback voltage V_(FB) from the output filter comprising inductor148 and capacitor 152.

In FIG. 1, the comparator mechanism 112 is constructed as a switchedcapacitor network with multiple phases of operation. In the context ofcircuit 100, during the idle phase, when the high side FET 104 is on,i.e., high side conduction period, V_(REF) is sampled and output bymultiplexer 116 at the input to capacitor 119. Also during the idlephase, switch 118 is closed so the output of capacitor 119 (and voltageinput to comparator 114) is held to ground. Thus, capacitor 119 storesthe charge corresponding to the magnitude of the reference voltage. Inthe next phase, when the low side conduction period is initiated, thatis, when the low side FET 108 is on, the switch 118 is opened, thereference voltage V_(REF) is essentially disconnected, and the capacitor119 continues to hold the stored charge of V_(REF). During the low sideconduction period, the multiplexer 116 outputs V_(FB) as the input tocapacitor 119, so the output of capacitor 119 now reflects thedifference of V_(FB)−V_(REF).

In FIG. 1, comparator mechanism 112 is referred to herein as a “sampleand hold comparator,” because the reference voltage V_(REF) is beingsampled by capacitor 119. That is, when the difference of V_(FB)−V_(REF)is calculated at the voltage input to comparator 114, V_(REF) is asampled value.

Configuring comparator mechanism 112 as described above creates aneffective threshold of (V_(FB)−V_(REF)) for static V_(REF) inputs at thecomparator mechanism 112. The third comparator mechanism input terminal,the current sensing input, is an input to comparator 114 and monitorsthe current I_(L) delivered through inductor 148 of the output filter.In one embodiment, as illustrated in FIG. 1, the voltage drop across thelow side FET 108, that is, between Vx node 132 and Gnd 136, is providedto the current sensing input as a measure proportional to inductorcurrent I_(L) across inductor 148. In this embodiment, the voltage dropacross the low side FET 108 will be equal to the current throughinductor 148, I_(L), as processed with, for example, multiplied by, theknown resistance across the source and drain of the low side FET 108(“R_(DSON)”). Other alternative embodiments providing a measure of thecurrent I_(L) delivered to the output filter are contemplated. In onealternative embodiment, a resistor is coupled to provide a measure ofthe current I_(L). For example, a resistor can be coupled in series withinductor 148, and the voltage across this resistor is provided to thecurrent sensing input of comparator 114. This additional resistor can bean external sense resistor coupled to the output filter, or the resistorcould be integrated as a component of the output filter or of the powerswitch, depending on the desired configuration. In these alternativeembodiments incorporating a resistor as the current sensing component,the resistance of the additional resistor is used in place of theR_(DSON) value in the calculations described herein. In anotheralternative embodiment, the voltage across inductor 148 is measured anddelivered to the current sensing input of comparator 114, with aneffective resistance value of inductor 148 substituted for the R_(DSON)value in the calculations below.

The comparator mechanism 112 will therefore trigger when the differencebetween the feedback voltage input and the reference voltage inputequals the voltage at the current sensing input, referred to herein as a“trigger event” or “triggering event,” as represented below:(V_(FB)−V_(REF))=−I _(L) ·R _(DSON)V_(FB)=V_(REF) −I _(L) ·R _(DSON)

In FIG. 1, the output of comparator mechanism 112 is coupled to a statemachine register 122, which is in turn coupled to a timer block 124. Thestate machine register 122 and timer 124 cooperate to control theswitching of high side switch 104 and low side switch 108. In oneembodiment, register 122 is an SR latch with timer 124 connected to the“R” reset input. The “Q” output of register 122 is coupled to both: (1)the select line of multiplexer 116, and (2) the switches 104 and 108.Because the output of register 122 is also coupled to the select line ofmultiplexer 116, the sampled input and output of the multiplexer iscontrolled by the same mechanism that causes the selection of the highside switch 104 or the low side switch 108. In addition, the output ofregister 122 is coupled to both: (3) switch 118, to synchronize theopening of switch 118 during the low side conduction period and closingof switch 118 during the high side conduction period, and (4) an inputto timer 124 to signal a reference time, a fixed time after which thetimer 124 will activate.

In FIG. 1, the output of comparator 114 goes from low to high when thetrigger event occurs. That is, when the low side FET 108 is on, and theinductor current signal I_(L) crosses the voltage error signalV_(FB)−V_(REF), the comparator 114 output goes to ‘1,’ causing theregister 122 to be set. When register 122 is set, in the example of anSR latch, the Q output goes high, turning on the high side FET 104.Responsive to the Q output going high, the timer 124 will activate the Rinput of register 122 a fixed time later to reset the latch back to ‘0’at the Q output and initiate the low side conduction period, i.e., turnon the low side FET 108. In this way, regulator 100 is provided with afixed high side (104) on time. The low side (108) on time, however,varies to provide the desired regulation.

The output of comparator 114 and, therefore, comparator mechanism 112,causes latch 122 to trigger a low to high FET switching transition athigh side and low side switches 104 and 108 when the error in theregulated output voltage (V_(FB)−V_(REF)) is equal to a proportionallyscaled value of the output inductor current, in the embodiment of FIG.1, I_(L)*R_(DSON). This method of control is known as current modecontrol since information regarding the regulator's output current isused to help establish the deterministic switching behavior of thevoltage regulator. Current mode control provides regulation of theoutput voltage using the output current through the output filter andthrough the load. Embodiments of the present invention that incorporatecurrent mode control protect the regulator 100 from delivering excessivecurrent and provide superior regulator performance by considering bothoutput current and output voltage in determining the cycle-to-cyclecontrol of the power FET state.

In particular, embodiments of the present invention incorporateprinciples of valley current mode control, which involves leaving thelow side FET 108 on until the output current I_(L) ramps down to asufficiently low value to trigger the end of the low side conductionperiod. In one embodiment, output voltage regulation is achieved bycombining this valley current mode control technique with fixed on timeof the power switch. That is, the high side FET 104 is switched on for afixed amount of time. In one example, the on time of the high side FET104 is set by a timer so that it does not change. With fixed on time,the actual on time of the high side FET 104 can be programmed or set asdesired, but is generally not controlled. In one embodiment, applyingprinciples of valley current mode control, however, the switching of lowside FET 108 is controlled. After the high side FET 104 is disabled, thelow side FET 108 is enabled and left to conduct until the combination ofthe output voltage error (from the reference) and the current sensefeedback indicates the low side FET should be disabled and the high sideFET should be re-enabled.

In an alternative embodiment of the present invention, the timer 124 ofFIG. 1 is replaced with clock having a constant frequency. In thisembodiment, the activation of the R input of register 122 to reset theregister and initiate the low side conduction period is controlled bythe steady clock signal. The frequency of the clock can be programmed orset as desired.

As mentioned above, in an embodiment implementing valley current modecontrol, the low side FET 108 is left on until the output current rampsdown to a sufficiently low value to trigger the end of the low sideconduction period. While embodiments of the invention are oftendescribed herein with regard to valley current mode control, othercontrol techniques such as such as peak current mode control orhysteretic current mode control may also be utilized. For instance, peakcurrent control can be employed when the voltage error signal isproportional to the intra-cycle peaks of the sensed current. This wouldgenerally involve the modulation being done during the high side FET 104conduction period, and the current sensing therefore being performedacross high side FET 104. In an embodiment based on hysteretic currentmode control, an additional comparator mechanism, constructed in similarfashion as comparator mechanism 112 or 200, can have a current sensinginput coupled at the output of high side FET 104 to respond at theappropriate time by switching from the high side conduction period tothe low side conduction period.

Other implementations incorporating aspects of the present inventionperform current sensing using other techniques that are not based on theFET 104 or FET 108 R_(DSON) value. For instance, current sensing can bebased on inductor DC resistance, explicit sense resistors, and otheractive circuitry measuring the current I_(L) being passed to inductor148 of the output filter.

As an alternative to regulator 100 depicted in FIG. 1, anotherembodiment of comparator circuitry can continuously track V_(REF) ratherthan use the sample and hold technique described above. Such a designmay be referred to as a “double differential” design, and a mechanism ofcomparator circuitry 200 shown in FIG. 2 replaces comparator mechanism112 of FIG. 1. In the embodiment of FIG. 2, four input capacitors 204,208, 212, and 216 replace the multiplexor 116, capacitor 119 and switch118 of comparator mechanism 112, while the remainder of the circuitry ofthe regulator would otherwise be like that as shown in FIG. 1 and asdescribed above. In this case, comparator mechanism 200 willcontinuously track V_(REF) during the comparison phase so that dynamicchanges of the reference voltage can be accommodated before the nextcomparison cycle.

In FIG. 2, the comparator mechanism 200 is utilized for differentialsensing of the current feedback (V_(x+)−V_(x−)) at switching node 132 ofFIG. 1 to account for variations in the relative grounding of thecurrent feedback and voltage feedback signals. A separate capacitor 204is coupled at the V_(x+) input to comparator 114, and another capacitoris coupled at the V_(x−) input to comparator 114. In FIG. 2, in oneembodiment based on valley current mode control, V_(x+) refers to node132 in FIG. 1 labeled Vx, and V_(x−) refers to node 136 connected to thesource of the low side FET 108 in FIG. 1. Thus, the current sensingdifferential voltage (V_(x+)−V_(x−)) is measured across low side FET108. In an alternative embodiment based on peak current mode control,the differential voltage across high side switch 104 is monitored; thus,V_(x+) refers to node 132 and V_(x−) refers to node 128, or V_(DD).Alternatively, as described above with reference to FIG. 1, the currentsensing differential voltage (V_(x+)−V_(x−)) can be sensed across theinductor 148 of the output filter or across a resistor, such as anexternal sense resistor, coupled in series with inductor 148. In FIG. 2,separate capacitors 212 and 216 are coupled at the V_(REF) and V_(FB)inputs, respectively, to provide for sensing a differential voltage atan input of comparator 114, as shown in FIG. 2. The comparator 114,therefore, is coupled to monitor the differentials between the voltageand current-sensed signals and trigger when the difference between thefeedback voltage input and the reference voltage input (V_(FB)−V_(REF))equals the voltage at the current sensing input (V_(x+)−V_(x−)).

FIGS. 3A and 3B are schematic diagrams of voltage regulators 300A and300B, respectively, constructed according to embodiments of theinvention.

Regulators 300A and 300B are similar to regulator 100 in many respects,with like reference numerals indicating like parts, but differ fromregulator 100 by the addition of alternative embodiments of a slow speedintegrator mechanism comprising an integrator to eliminate the finiteoutput impedance of the current mode voltage regulator 100. Theintegrator mechanism, described in greater detail below, adds anintegrative term with high gain to boost the overall gain back tonear-zero error, and does so with a slower response time.

In the embodiments of FIGS. 3A and 3B, the integrator mechanismcomprises integrator 142 in conjunction with a resistor 138. In FIG. 3A,one input terminal of the integrator 142 is coupled to the referencevoltage 144, and the other input terminal is connected to the feedbackvoltage 140. The output of integrator 142 is connected to the secondvoltage input of the multiplexer 116. The resistor 138 is coupled in thefeedback voltage path 140 at the second voltage input of the multiplexer116. FIG. 3B illustrates an alternative construction of the integratormechanism, in which the input terminals of the integrator 142 aresimilarly coupled to the reference voltage and the feedback voltage.However, in FIG. 3B, the second voltage input terminal of multiplexer116 remains directly connected to the feedback voltage 140, while theoutput of integrator 142 is coupled to the current sensing input ofcomparator mechanism 112. The resistor 138 is coupled between thecurrent sensing input of comparator mechanism 112 and the node Vx.

In FIGS. 3A and 3B, integrator 142 senses the feedback voltage andreference voltage and is configured to minimize the difference betweenthese sensed voltages. The integrator 142 outputs an integrativecorrection signal, a current in the example of FIGS. 3A and 3B, intoresistor 138 that causes a voltage drop that provides an adjusted signalto the comparator mechanism 112. In FIG. 3A, the adjusted signal isprovided at the second voltage input of multiplexer 116. In FIG. 3B, theadjusted signal is provided at the current sensing input of comparatormechanism 112. Both architectures of FIGS. 3A and 3B provide a staticoffset to the comparator mechanism 112.

Embodiments of the present invention as constructed in FIGS. 3A and 3Butilize the integrative elements 138 and 142 to inject a correction terminto the regulator architecture to account for the finite impedance ofthe current mode regulator. The correction term need not be implementedas shown in the depicted embodiments, as a correction term may beintroduced at any number of points in a regulating circuit (via anintegrator, etc.)

The circuits and methods described with reference to FIGS. 3A and 3B aretwo of many possible implementations for introducing an integrativecorrection signal into a voltage regulator. For example, in analternative embodiment to FIG. 3A, resistor 138 is coupled betweenV_(REF) and the integrator output rather than between V_(FB) and theintegrator output. In this way, the output of the integrator mechanismis coupled to the reference voltage input. In another alternativeembodiment, rather than connecting the integrator in a feedbackconfiguration as shown in FIGS. 3A and 3B, the integrator output couldbe coupled directly to comparator 114 so the integrative correctionsignal adjusts the threshold of comparator 114. This represents analternative to introducing the integrative correction signal into one ofthe input signals to the comparator, in the embodiments described above.

Regulators incorporating an integrator mechanism are capable ofoperating with zero static output impedance, in other words, without aninherent drop in output voltage as the load current increases (known asdroop). This allows incorporation of the regulator in larger systemswhere zero or minimal droop is specified. Because the integrator is notin a high speed feedback path, it can be implemented in smaller area andwith lower current consumption than designs incorporating a conventionalfeedback error amplifier approach. Also, since the integrator is onlyremoving the finite error due to current mode control and not performingthe high speed feedback signal processing and modulation, the integratorand the overall regulator can be designed in a relatively small area andwith limited supply current consumption.

In some of the embodiments described above, in which it is desirable tosense the output current I_(L) during the low side conduction period,current sensing is achieved by monitoring the voltage at switching nodeVx 132 at the output of the power switch. This current sensing voltage,in one embodiment, is the voltage across low side FET 108. In analternative embodiment, in which current sensing is performed during thehigh side conduction period, the current sensing voltage can be measuredacross high side FET 104. In either case, the current sensing voltage isgenerally proportional to the current I_(L) output through inductor 148with a scaling factor of the low side FET resistance during that phaseof operation, that is, when the low side FET is on. In an alternativeembodiment, in which current sensing is performed during the high sideconduction period, the current sensing voltage would be measured whenthe high side FET is on, with a scaling factor of the high side FETresistance.

One issue with measuring current by sensing the voltage across atransistor or other component is that the effective resistance of thetransistor, e.g., R_(DSON), is a factor. From lot-to-lot, and over thelifetime of production, the characteristics of a FET can vary. Thisincludes the resistance of the transistor, for instance, depending onwhen it is manufactured (“process” parameter). In addition, theresistance can change in response to temperature variations(“temperature” parameter), since a FET has a temperature coefficient forits resistance. The resistance can also change in response to differentsupply voltages—the resistance generally decreases as the supply voltageincreases (“voltage” parameter). Each of theseprocess-voltage-temperature (PVT) parameters contributes to fluctuationsin the resistance of the FET. Thus, in some embodiments in which PVTvariations could be an issue, it is desirable to sense the outputcurrent I_(L) in a manner that is independent of the resistance across aFET or other component at which the current is monitored.

In one embodiment, in FIG. 4, a resistance adjustment device 400 can beconstructed using integrated circuit fabrication techniques andincorporated as a component of the voltage regulators described above.The resistance adjustment device 400 can be coupled between the node atwhich the current sensing voltage is measured, node Vx 132 in thisexample, and the current sensing input(s) of comparator mechanism 112 orcomparator mechanism 200 of FIG. 2. As described in greater detailbelow, resistance adjustment device 400 is configured to enable currentsensing in a manner independent of the resistance associated with acomponent at which the current sensing voltage is measured, such as theR_(DSON) value of low side FET 108. Such a component is referred toherein as a “current sensing component.” A resistance adjustment device400 constructed in accordance with embodiments of the present inventionsenses the voltage across the current sensing component and performsoperations to effectively replace the resistance of the component with areference resistance, thus canceling out possible resistance variationsas described above.

FIG. 5 shows a diagram of one implementation of resistance adjustmentdevice 400 with a first input 504 a coupled to switching node 132 and asecond input 504 b coupled to terminal 136 of circuit 100. In this way,a differential voltage of I_(L)*R_(DSON), measured across low side FET108, is provided as an input to resistance adjustment device 400. Inthis example, the resistance of low side FET 108 is desired to beremoved from the calculations described herein to measure the outputcurrent I_(L). In other examples, when the current I_(L) is measuredacross another current sensing component, for instance, high side FET104, the inputs 504 a and 504 b of resistance adjustment device 400 canbe connected across that component, e.g., at V_(DD) node 128 and Vx node132 to remove the variation in its resistance from the current sensingcalculations described herein. Resistance adjustment device 400 furtherincludes output terminals 508 a and 508 b connected to the currentsensing input 408 of comparator mechanism 112, as shown in FIG. 4 or tothe differential current sensing inputs V_(x+) and V_(x−) of comparatormechanism 200, shown in FIG. 2.

In FIG. 5, resistance adjustment device 400 performs a transfer functionin which a factor of R_(REF)/R_(DSON) is applied to the input voltageprovided at terminals 504 a and 504 b. Thus, in one embodiment,resistance adjustment device 400 converts the sensed voltage ofI_(L)*R_(DSON) to I_(L)*R_(REF), a measure which is based on apredetermined reference resistance, rather than the potentially variableR_(DSON) value of the current sensing component, in this case, low sideFET 108. In this embodiment, the adjusted voltage I_(L)*R_(REF) isprovided to the current sensing input(s) of the comparator mechanism 112or 200 in place of the current sensing voltage measured across low sideFET 108. The reference resistance, R_(REF), is generally a controllableconstant, as described in greater detail below, thus providing a morestable current sensing voltage measurement across possible PVTvariations. That is, the scaled I_(L)*R_(REF) value can be PVTindependent. In some implementations, as further described below, theR_(REF)/R_(DSON) transfer function of resistance adjustment device 400effectively divides the current sensing voltage down to a smaller butdeterminable level.

FIG. 6 shows a diagram of a circuit 600 configured to sense the voltageacross a current sensing component, in this example, low side FET 108,in a manner that is independent of possible resistance variationsassociated with that switching component. The circuit 600 allows for thecurrent measured across a component having a resistance susceptible toPVT variations to be replicated with a determinable scaling factor,K_(I). The circuit 600 represents one implementation of a resistanceadjustment device 400 configured to replicate the current passingthrough the switching component of the power switch or other componentat which the output current I_(L) is desired to be measured.

In FIG. 6, the circuit 600 incorporates one or more matching components,which share PVT characteristics with the current sensing component atwhich I_(L) is measured. The matching component(s) can be identicallymatched or ratiometrically matched to the current sensing component. Inone implementation, as shown in FIG. 6, the matching component is a FEThaving similar physical characteristics as the low side FET of the powerswitch. For example, the matching FET may be sized relative to the lowside FET such that its resistance is a factor K_(I) times the resistanceof the low side FET. To realize such similarities, the components arepreferably built as part of the same integrated circuit fabricationprocess. For instance, if a matching FET is manufactured at the sametime as a FET of the power switch, they will often have the same PVTcharacteristics. In such contemporaneous fabrications, the matchingcomponent and the current sensing component will often share the sameprocess and temperature characteristics, because they are on the samedie, and they can be connected to the same voltage supply. This servesto compensate for any PVT fluctuations in the current sensing component,such as low side FET 108, as described in greater detail below.

In FIG. 6, in one embodiment, the circuit 600 uses a scaling factorK_(I) defined as the physical device size ratio between the currentsensing component and a matching component of the resistance adjustmentdevice. In the example of FIG. 6, the physical device size ratio, K_(I),is determined based on the gate width of the low side FET 108 inrelation to the gate width of a matching FET 612. The scaling factorK_(I) can be a large value in implementations where the physical areaoccupied by the current sensing component is large in relation to thearea occupied by the matching FET.

In the implementation of FIG. 6, matching FET 612 has an effectiveresistance of K_(I)*R_(DSON), that is, the scaling factor applied to theeffective resistance of low side FET 108. A current reference, I_(REF),is provided as an input to the drain of the matching FET 612. In someimplementations, this current reference is provided on-chip with theresistance adjustment device 400. Using appropriate integrated circuitdesign techniques, I_(REF) can be provided along with a referencevoltage, such as V_(REF) described above, as circuit componentscomprising an integrated circuit. In FIG. 6, at node 616, the voltage isI_(REF)*K_(I)*R_(DSON).

In FIG. 6, resistance adjustment device 400 incorporates a first voltagedivider unit comprising voltage dividing components 620 a and 620 b. Inone embodiment, the voltage dividing components 620 a and 620 b areimplemented as FETs having effective resistances R₁ and R₂,respectively. In alternative embodiments, other components havingeffective resistances can be substituted for FETs 620 a and 620 b shownin FIG. 6. The drain of FET 620 a is coupled to senseI_(REF)*K_(I)*R_(DSON) at node 616. The source of FET 620 a is coupledto the drain of FET 620 b, at node 622, while the source of FET 620 b isconnected to node 136, in this implementation, ground. In this way, thevoltage dividing components 620 a and 620 b are configured to divide thevoltage, I_(REF)*K_(I)*R_(DSON), sensed at node 616, across therespective resistances R₁ and R₂ of the individual components 620 a and620 b. The midpoint voltage at node 622 between voltage dividingcomponents 620 a and 620 b is a ratio of these resistances. Forinstance, in FIG. 6, the voltage sensed at midpoint node 622 is theinput voltage of I_(REF)*K_(I)*R_(DSON), sensed at node 616 in thisexample, multiplied by R₂/(R₁+R₂).

In FIG. 6, an amplifier such as op-amp 624 is implemented with afeedback configuration such that its output is coupled to the gate ofFET 620 b, and a first input to the amplifier is coupled to thereference voltage, V_(REF). The second input of op-amp 624 is coupled tothe midpoint node 622 of the first voltage divider unit. By beingconnected in this manner, the op-amp 624 will adjust its output so itstwo inputs are equal to one another. Thus, op-amp 624 is operativelycoupled to force the voltage at node 622 to the V_(REF) value. Inparticular, by being coupled to the gate of FET 620 b, op-amp 624 willdrive the gate voltage and thereby adjust the R₂ value so the mid-pointvoltage at node 622 adjusts to the V_(REF) value. The resistance ratioof the voltage dividing components 620 a and 620 b can thus becalculated as:

$\frac{R_{2}}{R_{1} + R_{2}} = \frac{V_{REF}}{I_{REF} \cdot K_{I} \cdot R_{DSON}}$

In other words, the behavior of the first voltage divider unit isgoverned by the ratio of V_(REF) to I_(REF)*K_(I)*R_(DSON). To achievethis, the amplifier 624 is operatively coupled as described above toessentially adjust the R₂ value of FET 620 b, and thus affect theR₂/(R₁+R₂) value, so that V_(REF) is maintained at the midpoint node 622of the first voltage divider unit.

In alternative embodiments, it is possible to adjust both resistances(R₁ and R₂) or just R₁. For instance, when a wider variation in theR_(DSON) value of FET 108 is expected, it could be desirable to controlboth R₁ and R₂ to allow the circuit to operate over a wider range.Various implementations are contemplated to adjust the ratio of R₁ andR₂ to achieve the desired voltage divider ratio. Either R₁ or R₂, orboth resistances, can be adjusted as desired. Also, various feedback andadjustment circuit topologies can be implemented to actively control avoltage divider unit tuned to attenuate the voltageI_(REF)*K_(I)*R_(DSON) to V_(REF), with a matching voltage divider unitcoupled to sense the V_(x) voltage, as further described below.

In FIG. 6, the circuit 600 further includes a matching second voltagedivider unit, having voltage dividing components corresponding to thecomponents of the first voltage divider unit described above. In oneembodiment, the matching voltage divider unit includes voltage dividingcomponents 628 a and 628 b implemented, in this example, as FETs havingeffective resistances R₁ and R₂, respectively. Thus, the effectiveresistance of FET 628 a substantially matches that of FET 620 a, and theeffective resistance of FET 628 b substantially matches that of FET 620b. In alternative embodiments, other components having matchingeffective resistances can be substituted for these pairs of FETsimplemented in the respective voltage divider units.

In FIG. 6, the resistances R₁ and R₂ of the FETs comprising both thefirst and second voltage divider units can be independently controlledaccording to the gate drives at the respective FETs. Thus, for instance,the voltage input to the gate of FET 628 a will affect its R₁ value. Inone implementation of the circuit of FIG. 6, the same voltage, V_(DD),is provided to the gates of FETs 620 a and 628 a, ensuring that bothFETs have substantially the same R₁ value. Similarly, the gates of FETs620 b and 628 b are coupled to one another so that amplifier 624 drivesFET 628 b in the same manner as FET 620 b, described above. Othercontrol voltages can be provided to the gates of the matching pairs ofFETs in other implementations.

In the embodiment of FIG. 6, the drain of FET 628 a is coupled toswitching node 132, at which the output current I_(L) is sensed. Thesource of FET 628 a is coupled to the drain of FET 628 b, while thesource of FET 628 b is connected to terminal 504 b shared by the sourcesof low side FET 108 and FET 620 b. In this way, the voltage dividingcomponents 628 a and 628 b of the matching voltage divider unit areconfigured to divide the current sensing voltage at node 132 across therespective resistances R₁ and R₂ of the individual components 628 a and628 b.

In FIG. 6, because the first and second voltage divider units havesubstantially matching voltage dividing components and the same gatevoltages, the units generally exhibit the same voltage dividingcharacteristics according to the R₂/(R₁+R₂) ratio. For instance, in FIG.6, the voltage sensed at output terminals 508 a and 508 b (V_(SENSE)) ofthis implementation of resistance adjustment device 400 is the currentsensing voltage, I_(L)*R_(DSON) in this example, multiplied byR₂/(R₁+R₂). Because the R₂/(R₁+R₂) value was determined above asV_(REF)/(I_(REF)*K_(I)*R_(DSON)), using the first voltage divider unit,V_(SENSE) can be computed as follows:

$\begin{matrix}{V_{SENSE} = {I_{L} \cdot R_{DSON} \cdot \frac{R_{2}}{R_{1} + R_{2}}}} \\{= {I_{L} \cdot R_{DSON} \cdot \frac{V_{REF}}{I_{REF} \cdot K_{I} \cdot R_{DSON}}}} \\{= {I_{L} \cdot \frac{1}{K_{I}} \cdot \frac{V_{REF}}{I_{REF}}}}\end{matrix}$

Thus, the replacement current sensing voltage provided to the currentsensing input(s) of comparator mechanism 112 or comparator mechanism 200is (I_(L)*V_(REF))/(I_(REF)*K_(I)), with R_(DSON) having been replacedby the on-chip voltage reference (V_(REF)) divided by an on-chip currentreference (I_(REF)) and an area-based ratio of the current sensingcomponent to the matching component (K_(I)). Returning to FIG. 5, inthis implementation, the reference resistance, R_(REF), is the valueV_(REF)/(I_(REF)*K_(I)).

In the embodiment of FIG. 6, the circuit 600 is configured to leverage:a PVT independent voltage reference, a PVT independent currentreference, and a matching FET with dimensions K_(I) times smaller thanthe current sensing component which the matching FET is intended toreplicate. In one implementation, the circuit 600 incorporates anamplifier with matching voltage divider units and modulates the gatedrive and, hence, effective resistance, of one or more components in thefirst voltage divider unit so a node between voltage dividing componentsof the unit equals the voltage reference. The second voltage dividerunit is operatively coupled to divide down the I_(L)*R_(DSON) value, andthe corresponding output is proportional to I_(L) but independent ofR_(DSON).

In alternative embodiments to that described above with reference toFIG. 6, current sensing is performed during the high side conductionperiod, in which case the current sensing voltage can be measured whenthe high side FET 104 is on. In these alternative embodiments,principles described above are still applicable. In one embodiment, whenhigh side FET 104 is a p-channel FET rather than an n-channel FET,circuit 600 can be re-configured, as will be appreciated by thoseskilled in the art, including replacing the re-channel FETs 612, 620 a,620 b, 628 a, and 628 b, of FIG. 6 with appropriately interconnectedp-channel FETs. In another alternative embodiment based on high sidecurrent sensing, high side FET 104 can remain an n-channel device, andthe FETs comprising circuit 600 can remain as n-channel FETs configuredsubstantially the same as shown in FIG. 6. However, in this alternativeembodiment, input terminals 504 a and 504 b of the resistance adjustmentdevice can be coupled across V_(DD) and node Vx 132 rather than beingcoupled across Vx 132 and ground. Thus, in this alternative embodiment,Vx 132 can be viewed as serving as a virtual ground.

The embodiments described above with reference to FIGS. 5 and 6 providean essentially passive voltage divider on the current sensing voltagefor accurate current sensing, by converting the sensed voltage to aPVT-independent measure based on a predetermined reference resistancerather than the potentially variable R_(DSON) value of the currentsensing component. In FIGS. 5 and 6, the conversion is achieved withoutcoupling active components in the signal path between the node at whichthe current sensing voltage is measured, node Vx 132 in theseembodiments, and the current sensing input(s) of comparator mechanism112 or comparator mechanism 200 of FIG. 2. This signal path is referredto herein as the “current sensing signal path.”

In one embodiment, as shown in FIG. 6, the current sensing signal pathof circuit 600 has an input at node Vx 132, runs from node 132 to nodes504 a and 504 b, and from nodes 504 a and 504 b to an output atV_(SENSE) nodes 508 a and 508 b. Only passive components, namely voltagedividing components 628 a and 628 b of the second voltage divider unit,are coupled in this signal path. While components 628 a and 628 b aretransistors, in one embodiment, these components are configured tooperate essentially as resistors to divide down the voltage in thesecond voltage divider unit. Any active components, namely the amplifier624 coupled to the gate of FET 620 b and FET 628 b, are not coupled inthe current sensing signal path. The matching FET 612 and currentreference, I_(REF), are also not coupled in the current sensing signalpath.

Because there are no active components coupled in the current sensingsignal path, the speed of the passive voltage divider network can begoverned by parasitic elements and have fast response times. Theamplifier 624 in the embodiments above does not add any appreciabledelay to circuit 600. The amplifier and any associated circuitry canhave slower speeds, require smaller area, consume lower supply currents,and be easier to implement than conventional designs, since thecircuitry is not in the current sensing signal path.

Depending on the desired implementation, different circuitcomponents/mechanisms described herein can be fabricated so that theyshare the same substrate, e.g., are on the same die or chip. In analternative implementation, such circuit components and mechanisms canbe fabricated on different substrates, e.g., on different chips. Ineither implementation, such circuit components and mechanisms can beprovided in the same or different packages. For instance, a comparatormechanism fabricated on a first die could be interconnected with a powerswitch fabricated on a different second die, interconnected with oneanother as described above, and provided in the same package. In anotherexample, the comparator mechanism, integrator mechanism, or voltagedivider unit(s) could be implemented in a discrete controller separatefrom other circuit components in the embodiments described herein.

While the invention has been particularly shown and described withreference to specific embodiments thereof, it will be understood bythose skilled in the art that changes in the form and details of thedisclosed embodiments may be made without departing from the spirit orscope of the invention. The present invention should of course, not belimited to the depicted embodiments. In addition, although variousadvantages and aspects of the present invention have been discussedherein with reference to various embodiments, it will be understood thatthe scope of the invention should not be limited by reference to suchadvantages and aspects. Rather, the scope of the invention should bedetermined with reference to the appended claims.

What is claimed is:
 1. A voltage regulator comprising: a power switch having an input and an output, the power switch configured to provide a first voltage during a first conduction period and a second voltage during a second conduction period; an output filter coupled between the power switch output and an output terminal to be coupled to a load; and a comparator mechanism having a reference input coupled to a reference voltage, a feedback input coupled to sense a feedback voltage at the output filter, a current sensing input coupled to sense a current sensing voltage corresponding to a current provided to the output filter, and an output in communication with the power switch input, the comparator mechanism configured to trigger responsive to a difference between the feedback voltage and the reference voltage equaling the current sensing voltage, the triggering causing a transition of the power switch from the second conduction period to the first conduction period.
 2. The voltage regulator of claim 1, the first voltage being a high voltage, and the second voltage being a low voltage.
 3. The voltage regulator of claim 1, the first voltage being a low voltage, and the second voltage being a high voltage.
 4. The voltage regulator of claim 1, the current sensing voltage representing the current provided to the output filter multiplied by an effective resistance of a component of the power switch.
 5. The voltage regulator of claim 4, the power switch component comprising a low side transistor.
 6. The voltage regulator of claim 4, the power switch component comprising a high side transistor.
 7. The voltage regulator of claim 1, the current sensing voltage representing the current provided to the output filter multiplied by an effective resistance of a component of the output filter.
 8. The voltage regulator of claim 7, the output filter component comprising an inductor.
 9. The voltage regulator of claim 1, the current sensing voltage representing the current provided to the output filter multiplied by a resistance of a resistor coupled at the output filter.
 10. The voltage regulator of claim 9, the resistor being an external sense resistor.
 11. The voltage regulator of claim 1, the comparator mechanism comprising a sampling circuit coupled to sample the reference voltage during the first conduction period.
 12. The voltage regulator of claim 11, the sampling circuit of the comparator mechanism comprising: a multiplexer having the reference input and the feedback input as inputs, a capacitor having an input coupled to an output of the multiplexer, a switch coupled to an output of the capacitor, and a comparator having a first input coupled to the capacitor output, the current sensing input as a second input, and an output representing the comparator mechanism output, the comparator output in communication with a select line of the multiplexer and a control of the switch.
 13. The voltage regulator of claim 12 further comprising: a state register having an input coupled to the comparator output and an output coupled to: the power switch input, the multiplexer select line, and the switch control, and a switching control unit coupled to a control input of the state register, the state register capable of: initiating the transition of the power switch to the first conduction period responsive to the comparator mechanism output, and initiating a transition of the power switch to the second conduction period responsive to the switching control unit, including causing the multiplexer to output the reference voltage during the first conduction period and the feedback voltage during the second conduction period, and causing the switch to be closed during the first conduction period and open during the second conduction period.
 14. The voltage regulator of claim 13, the switching control unit comprising a timer.
 15. The voltage regulator of claim 14, the timer having a fixed on time.
 16. The voltage regulator of claim 14, the timer having a variable on time.
 17. The voltage regulator of claim 1 further comprising: a state register having an input coupled to the comparator mechanism output and an output coupled to the power switch input, and a switching control unit coupled to a control input of the state register, the state register capable of: initiating the transition of the power switch to the first conduction period responsive to the comparator mechanism output, and initiating a transition of the power switch to the second conduction period responsive to the switching control unit.
 18. The voltage regulator of claim 17, the switching control unit comprising a timer.
 19. The voltage regulator of claim 17, the switching control unit comprising a clock.
 20. The voltage regulator of claim 1, the comparator mechanism comprising a continuous tracking circuit coupled to sense the reference voltage.
 21. The voltage regulator of claim 20, the continuous tracking circuit of the comparator mechanism comprising: a first current sensing voltage input and a second current sensing voltage input coupled to sense a differential voltage representing the current sensing voltage.
 22. The voltage regulator of claim 21, the first current sensing voltage input and the second current sensing voltage input of the continuous tracking circuit coupled to sense the differential voltage across a component of the power switch.
 23. The voltage regulator of claim 22, the power switch component comprising a transistor.
 24. The voltage regulator of claim 21, the first current sensing voltage input and the second current sensing voltage input of the continuous tracking circuit coupled to sense the differential voltage across a component of the output filter.
 25. The voltage regulator of claim 24, the output filter component comprising an inductor.
 26. The voltage regulator of claim 24, the first current sensing voltage input and the second current sensing voltage input of the continuous tracking circuit coupled to sense the differential voltage across a resistor coupled at the output filter.
 27. The voltage regulator of claim 26, the resistor being an external sense resistor.
 28. The voltage regulator of claim 21, the first current sensing voltage input of the continuous tracking circuit comprising a capacitor.
 29. The voltage regulator of claim 21, the second current sensing voltage input of the continuous tracking circuit comprising a capacitor.
 30. The voltage regulator of claim 21, the reference input of the comparator mechanism comprising a capacitor.
 31. The voltage regulator of claim 21, the feedback input of the comparator mechanism comprising a capacitor.
 32. The voltage regulator of claim 21, the continuous tracking circuit of the comparator mechanism comprising: a comparator having a first input coupled to the reference input and the feedback input, and a second input coupled to the first current sensing voltage input and the second current sensing voltage input, and an output representing the comparator mechanism output.
 33. The voltage regulator of claim 1 further comprising: an integrator mechanism coupled to sense the feedback voltage and the reference voltage and having an output coupled to one of the comparator mechanism inputs, the integrator mechanism configured to determine a difference between the feedback voltage and the reference voltage and output an adjusted signal, based on the determined difference, to the comparator mechanism.
 34. The voltage regulator of claim 33, the integrator mechanism comprising an integrator and a resistor.
 35. The voltage regulator of claim 33, the integrator mechanism output coupled to the feedback input of the comparator mechanism.
 36. The voltage regulator of claim 33, the integrator mechanism output coupled to the current sensing input of the comparator mechanism.
 37. The voltage regulator of claim 33, the integrator mechanism output coupled to the reference voltage.
 38. The voltage regulator of claim 33, the integrator mechanism configured to adjust a comparator threshold of the comparator mechanism.
 39. The voltage regulator of claim 1, the power switch comprising: a first transistor coupled between the first voltage and the power switch output.
 40. The voltage regulator of claim 39, the power switch further comprising: a second transistor coupled between the second voltage and the power switch output.
 41. The voltage regulator of claim 39, the power switch further comprising: a diode coupled between the second voltage and the power switch output.
 42. The voltage regulator of claim 1, the comparator mechanism and the power switch situated on a common substrate.
 43. The voltage regulator of claim 1, the comparator mechanism and the power switch situated on different substrates.
 44. The voltage regulator of claim 1, the comparator mechanism situated in a discrete controller separate from the power switch.
 45. A voltage regulation method comprising: providing a first voltage during a first conduction period and a second voltage during a second conduction period; sensing a current sensing voltage corresponding to a current provided to an output filter; and determining a trigger event using a comparator mechanism when a difference between a feedback voltage at the output filter and a reference voltage equals the current sensing voltage, the trigger event causing a transition from the second conduction period to the first conduction period, the comparator mechanism having a reference input coupled to the reference voltage, a feedback input coupled to sense the feedback voltage, a current sensing input coupled to sense the current sensing voltage, and an output capable of outputting a signal indicative of the trigger event.
 46. The method of claim 45, the first conduction period being a high conduction period.
 47. The method of claim 45, the first conduction period being a low conduction period.
 48. The method of claim 45, the method further comprising: sampling the reference voltage during the first conduction period.
 49. The method of claim 45, the method further comprising: continuously sensing the reference voltage.
 50. The method of claim 45, the method further comprising: determining a difference between the feedback voltage and the reference voltage; and providing an adjusted signal, based on the determined difference, to one of the comparator mechanism inputs. 